The present invention relates to a high frequency filter, in particular, relates to a novel filter of dielectric waveguide type, which is suitable for use especially in the range from the VHF bands to the comparatively low frequency microwave bands.
First, three prior filters for the use of said frequency bands will be described.
FIG. 1 shows the perspective view of a conventional interdigital filter, which has been widely utilized in the VHF bands and the low frequency microwave bands. In the figure, the reference numerals 1-1 through 1-5 are resonating rods which are made of conductive material, 2-1 through 2-4 are gaps between adjacent resonating rods, and 3 is a case. 3-1 through 3-3 are conductive walls of said case 3. A cover 3-4 of the case 3 is not shown for the sake of the simplicity of the drawing. A pair of exciting antennas 4 are provided for the connection of the filter to an external circuit. The length of each of the illustrated resonating rods 1-1 through 1-5 is selected as to be substantially equivalent to one quarter of a wavelength, and one end of the resonating rods are short-circuited alternately to the confronting conductive walls 3-1 and 3-2, while the opposite ends thereof are free standing.
However, said interdigital filter has the disadvantage that each of the resonating rods is fixed alternately to the confronting two conductive walls, in order to obtain the enough coupling coefficient between each resonating rods, and so, the manufacture of the filter is cumbersome and subsequently the filter is costly. If each of the resonating rods were mounted on a single wall, the coupling between each of the resonating rods would not be enough and the characteristics of a filter would not be satisfactory.
Now, the theoretical analysis that the coupling between each of the resonators would be insufficient if the resonators were arranged in line on the single conductive side wall, will be described below in accordance with FIGS. 2(A) through 2(C) and FIG. 3.
In realizing a high frequency filter with an excellent electric characteristic, it is very important how to build up coupling between adjacent resonators. More specifically, however high Q, or a small loss the resonators or, a loss in the coupling means between resonators results in an increase in the filter loss. Accordingly, it has been practice to provide the coupling between resonating rods by air gaps made by suitably spacing the resonating rods as shown in FIG. 1. However, if the resonating rods should be fixed to the single bottom surface 3-1, the coupling between the adjacent resonators would be very small, and a filter with a desired band width could not be obtained.
In FIGS. 2(A) through 2(C), solid line arrows and dotted line arrows represent vectors of electric field and magnetic field of high frequency, respectively. FIG. 2(A) is a horizontal sectional view of FIG. 1 on the conditions that one of the ends of the resonating rods 1-1 and 1-2 are short-circuited to the single conductive bottom surface 3-1, and FIGS. 2(B) and 2(C) are vertical sectional views. In the figures, 3-3 and 3-4 show upper and lower bottom surfaces, as in the case of FIG. 1.
Now, the coupling between the resonating rods 1-1 and 1-2 will be analyzed by separately taking the magnetic coupling and the electric coupling. It should be noted that the electric field and the magnetic field in FIGS. 2(A) through 2(C) are TEM mode.
Concerning the magnetic coupling, .phi..sub.1 is the high-frequency magnetic flux around the resonating rod 1-1, and I.sub.1.phi. is a high-frequency current accompanied by said flux .phi..sub.1. The directions of .phi..sub.1 and I.sub.1.phi. are as shown in the figures. The flux .phi..sub.2 induced around the resonating rod 1-2 by the flux .phi..sub.1 can have two directions. The first direction is shown in FIG. 2(A) wherein .phi..sub.1 and .phi..sub.2 cancel each other in the gap 2-1, resulting that the flux of .phi.=.phi..sub.1 =.phi..sub.2 surrounds both the resonating rods 1-1 and 1-2 as shown in FIG. 2(B). In this case it should be noted that an electric current I.sub.2.phi. flows in the resonating rod 1-2 in the direction as shown in the drawing, due to the flux .phi.. Thus, the magnetic coupling is performed as shown in FIG. 2(B) with the coupling coefficient k.sub..phi.. The second direction of .phi. which is induced on the resonating rod 1-2 by the flux .phi..sub.1 is the case that the flux .phi..sub.2 is in the opposite direction of FIG. 2(A), and in this case, both the fluxes .phi..sub.1 and .phi..sub.2 exist in the gap 2-1 as shown in FIG. 2(C), and there is no coupling between .phi..sub.1 and .phi..sub.2 in case of FIG. 2(C).
Then, the electric field coupling will be analyzed. E.sub.1 is the high-frequency electric field emanating from the surface of the resonating rod 1-1, and I.sub.1E is a high-frequency electric current accompanied by the electric field E.sub.1. The directions of E.sub.1 and I.sub.1E are shown in the figures. The electric field E.sub.2 induced on the surface of the resonating rod 1-2 by the electric field E.sub.1 can have two directions. The first direction is shown in FIG. 2(A) wherein E.sub.1 and E.sub.2 are mutually continuous in the gap 2-1, resulting in that the electric field E=E.sub.1 =E.sub.2 surrounds both of the resonating rods 1-1 and 1-2 as shown in FIG. 2(C). In this case, it should be noted that an electric current I.sub.2E flows in the resonating rod 1-2 in the direction as shown in the figure, due to the electric field E. Thus, the electrical coupling is accomplished as shown in FIG. 2(C) with the coupling coefficient k.sub.E. The second direction of E.sub.2 induced on the resonating rod 1-2 by the electric field E.sub.1 is the case that the field E.sub.2 is in the opposite direction of FIG. 2(A), and in this case, there exists an electric field as shown in FIG. 2(B), and there is no coupling between the electric fields E.sub.1 and E.sub.2.
The aforesaid four combinations are not mutually independent, due to the nature of the electromagnetic field, and can be summarized into two quantities, namely, the magnetic field coupling k.sub..phi. shown in FIG. 2(B) and the electric field coupling k.sub.E shown in FIG. 2(C).
Now, attention is paid to the direction of currents in FIG. 2(A). More particularly, the directions of I.sub.1.phi. and I.sub.1E are the same with each other, and the direction of I.sub.2.phi. is opposite to that of I.sub.2E. Accordingly, the amount of the coupling k.sub.12 between the resonating rods 1-1 and 1-2 can be expressed by; EQU k.sub.12 =.vertline.k.sub..phi. -k.sub.E .vertline. (1)
Thus, the relations among k.sub.12, k.sub..phi. and k.sub.E can be defined by the formula (1). The variation of k.sub.12 with the distance (x) between the resonating rods 1-1 and 1-2 is shown in FIG. 3. This is due to the fact that both k.sub..phi. and k.sub.E monotonously decreases with the distance (x) on the principle of electromagnetics. However, since the coupling between resonators in FIGS. 2(A) through 2(C) is accomplished by TEM mode (Transverse Electric Magnetic mode), the absolute value of the coupling coefficient is very small, and further, since the coupling coefficient k.sub.12 decreases with the distance (x), said distance (x) must be very small for obtaining a sufficient coupling coefficient for a practical filter. However, in an actual filter, said distance (x) can not be small enough to provide the sufficient coupling coefficient, and so a filter in which resonators are arranged on a single conductive wall can not be embodied, instead, resonators have been arranged interdigitally as shown in FIG. 1.
FIG. 4 shows the perspective view of another conventional filter, which is a comb-line type filter, and has been utilized in the VHF bands and the low frequency microwave bands. In the figure, the reference numerals 11-1 through 11-5 are conductive resonating rods with one of the ends thereof left free standing while opposite ends thereof are short-circuited to the conductive wall 13-1 of a conductive case 13. The length of each resonating rod 11-1 through 11-5 is selected to be a little shorter than a quarter of a wavelength. The resonating rod acts as inductance (L), and capacitance (C) is provided at the head of each resonating rod for providing the resonating condition. In the embodiment, said capacitance is accomplished by the disks 11a-1 through 11a-5 and the conductive bottom wall 13-2 of the case 13. The gaps 12-1 through 12-4 between each of the resonating rods provides the necessary coupling between each of the resonating rods. A pair of antennas 14 are provided for the connection between the filter and external circuits.
With this type of filter, the resonating rods 11-1 through 11-5 are fixed on the single bottom wall 13-1 and the manufacturing cost can be reduced as far as this point is concerned, but there is the shortcoming in that the manufacture of the capacitance (C) with an accuracy of, for instance, several %, is rather difficult, resulting in no cost merit. Therefore, the advantage of a comb-line type filter is merely that it can be made smaller than an interdigital filter.
FIG. 5 shows a perspective view of a conventional dielectric filter. In the figure, 21-1 through 21-5 are dielectric resonators each of which has a suitable thickness with the cross sectional dimensions usually selected for satisfying resonating conditions, while the length of each resonator is determined by considering such factors as unloaded Q.sub.u, and/or a spurious characteristics. The resonators 21-1 through 21-5 are fixed on a dielectric plate 23-1 which has a small dielectric constant and placed in a shielding case 23. The gaps 22-1 through 22-4 are provided between the resonators in order to achieve the desired degree of coupling between adjacent resonators. Also, a pair of exciting antennas 24 are provided for the coupling of the filter with an external circuit.
However, this type of filter has the shortcoming in that the size of each resonator is rather large even when the dielectric constant of the material of the resonators is as large as possible. Therefore, it is hardly practical for actual application of this filter in the VHF bands and the low frequency microwave bands.